31 Januaray 2015

New Tube: Nutube 6P1

Back to the Future: Nutube 6P1
Special thanks to long-time reader Larry Owens for tipping me off about this new, new development. What new development? A new, modern, small, long-life, low-power-consumption triode: the 6N1, made by Noratake for Korg. See Synthtopia and Sonicstate for more details. This is the future development that I have been waiting for a long, long time. About 25 years ago, I read that the Japanese TV makers, such as Sony and NEC, had developed new cathodes for their CRTs, which didn't wear away in a few thousand hours the way a typical triode's cathode does. Well, I instantly thought, why not make new audio tubes that used that new cathode technology? Hell, why not make laser cut grids out of metal foils? Why not use ceramic support structures within the tube? Why not use robots, rather than people, to build high-precision tubes? The list goes on and on and on...

Imagine if the vacuum tube had never been invented, that we all grew up with FETs and transistors, and that just now some clever research scientist figured out how to make a vacuum-tube triode. Would it look anything like a 12AX7? I doubt it. Instead, it would look very much like the 6P1, compact and precise.

The only thing that troubled me in reading about the 6P1 was its intended market, the music-amplifier market, which although I acknowledge is vastly larger than the puny high-end audio market. So? I fear that they might have designed the new 6P1 triode to be less linear than they could have, striving for a riper, more musical sound. Mind you, I have nothing against a more musical sound—as long as it comes from musical instruments and music halls, not playback equipment. This might seem an odd thing for someone who has designed harmonic restoration circuits to say, but those circuits were meant to correct and undo other defects in the signal chain. Of course, if your tastes dictate always a riper sonic overlay on all signals, great, go for it. De gustibus non est disputandum: in matters of taste, there can be no disputes. I only wear sunglasses when the sun is too bright; others, all the dang time—even at night. To each his own.

By the way, long ago, they came up with the gridistor, which was a FET that yielded triode-like curves. This made sense, as the gridistor's internal structure was very much like a triode's, albeit in solid-state, not vacuum voided.

 

Even More

Even More Circlotron Circuits
The Circlotron offers one great advantage over the typical totem-pole arrangement of output devices: the same type of device can be used throughout and both can see identical, although differing in phase, input signals. NPN and PNP output transistors never exactly match each other—nor do N-channel and P-channel MOSFETs. Of course, even with two of the same devices, say either two KT88 tubes or two 2N3055 NPN transistors, mismatches are not only possible, but common. Still, with some careful matching, it is possible to get a true matched pair. Why would that be important? Imagine that the left tires on your car were slightly smaller than the right tires. You could drive straight, with little effort. But driving through a tortuous road through the mountains could prove a torturous experience with mismatched tires. Well, with reproducing music, the same sort of problem arises, especially if the amplifier is a feedback-free design.

We can build totem-pole output stages with identical devices, either tube-based or transistor-based or MOSFET-based, but then we face the problem of delivering asymmetrical drive signals to the output devices.

In addition, feedback must be used in creating the asymmetrical drive signals or the output impedances from the two output devices will not match each other. For example, in the above circuit, the top triode functions as a cathode follower, which exhibits a low output impedance; in contrast, the bottom triode functions as a grounded-cathode amplifier, which presents a high output impedance. Feedback from the output to the bottom triode creates a cathode-follower-like performance from the bottom tube.

Well, if the Circlotron output stage offers so great an advantage, why don't solid-state amplifiers use it? A few do, but not many. Why not? We can amend Nietzsche's quip that

"One will seldom go wrong if one attributes extreme actions to vanity, moderate ones to habit, and petty ones to fear"

(#74 from his book, Human, All Too Human) with

"and sub-optimal engineering ones to increased expense."

Two floating power supplies are almost twice as expensive as a single, grounded bipolar power supply, and much cheaper than the four floating power supplies needed for a stereo Circlotron power amplifier.

Well, if we bite our wallet and pay the extra expense that a Circlotron output stage demands, will we have arrived at perfection? No. True enough that we have made a move in the right direction, but we do not necessarily arrive? Why not? Few are willing to run their Circlotron output stages, in spite of what their ad copy says, in true, honest-to-God class-A. Why not? Heat. Lots and lots of heat. Moreover, we will need huge arrays of output tubes (or transistors) and massive power transformers and heavy heatsinks and strong back to the lift the amplifier. In a world that is rapidly going class-D, class-A is a hard sell to make. (See blog number 284 for more information on class-A operation.) Thus, 99.999% of power amplifiers run in class-AB, which offers less distortion than class-B and less wasted heat dissipation than class-A. In other words, a compromise.

Well, couldn't a class-AB, push-pull, power amplifier that used two perfectly matched output devices present the same low distortion of a class-A amplifier? No, sadly. Why not? Imagine not only perfectly matched output devices, but perfectly linear output devices, devices so linear that they made rulers look bent out of shape.

Yet these perfectly-linear devices, when paired up and placed within a class-AB output stage, must distort.

In the class-A range of operation, the two devices work effectively in parallel, offering half of the output impedance and twice the transconductance of a single device. Once the output leaves class-A, however, and moves into class-B, with only one output device conducting, the output impedance doubles and the transconductance halves. So, while these perfectly-linear output devices would perfectly suit a class-A power amplifier, they would be ill suited to a typical class-AB output stage. (I remember reading in an issue of an RCA internal magazine that triodes, like the 2A3 and RCA 50 were best suited to class-A, push-pull use; whereas pentodes, like the 6L6 and 6V6, were best suited to class-AB, push-pull use. Why? The triode were more linear, while the pentodes exhibited a varying transconductance, making them a better match class-AB, push-pull operation.)

Effectively, while operating in class-A, the load appears twice as large in impedance, so an 8-ohm speaker appears as a 16-ohm load. How's that possible? Once one output device cuts off, the load is seen for what it really is, an 8-ohm load. Don't get this? Imagine a push-pull output stage that runs an idle current of 1A. Now, imagine an input voltage of +8V and a load resistance of 8 ohms. The top output device will increase its conductance to 1.5A, while the bottom device will decrease its to 0.5A, with the load seeing 1A of current flow, as 8V will develop across it. In other words, the +8V input signal only provoked a 500mA increase in current flow through the top device, which would imply a 16-ohm load resistance with a single output device. But in a push-pull output stage, the load sees the delta, the difference in conduction between the two output devices, so the 8-ohm load sees +500mA - -500mA, which equals 1A.

Once the push-pull output stage leaves the class-A overlap of conduction, the load only sees the delta in the single conducting device, so the load impedance is no longer effectively doubled. In this example, when the output voltage exceeds 16V. For example, if we increase the input signal voltage from 16V to 24V, the output current will go from 2A to 3A, the 1A increase implying an 8-ohm load, not a 16-ohm one. This break in continuity creates a glitch in the otherwise smooth transition, resulting in gm-doubling distortion, or what it should be called, gm-halving distortion, as in the overlap region things are fine.

Now, imagine that we could buy output devices that presented two transconductances, the first being exactly half of what the second is.

Such a device would make a terrible single-ended power amplifier. Much like a sharp jab in your sides from a pointed elbow, the twist in transconductance would thwack your ear.* On the other hand, such a device in a push-pull amplifier could—under the just right operating conditions—please and soothe your ears, just as your hand likes to run over new, taut, tidy, and smooth satin sheets. The procedure would be to set the idle current at exactly half of the transition current.

At exactly half of the transition current, the two output devices would perfectly overlap, creating a constant-gm transfer curve—or rather transfer line, as the line would not curve at all, as blue trace shows below.

What is happening is that in the class-A overlap of output-device conduction, the two half transconductances combine to equal second twice as large amount of transconductance. In other words, we still have gm doubling, but the doubling is against one half which results in unity. Mind you, if the output devices do exhibit two transconductances, but with the first one being less than the half of the second, then this trick will not work. Nor would the trick work if the second transconductance was less than the first transconductance.

Well, since we cannot buy these desirable output devices, being stuck with EL34, KT88, and BUZ901, IRF230, and MJ21194 output devices, what can we do? The usual approach is to employ gobs and gobs of negative feedback, so that the output is forced inline. Another far more subtle approach would be to apply a complementary pre-distorted drive signal, which would result in a linear transfer curve. Or, we could find a way to make the output stage behave as if it offered a constant transconductance.

 

 

Class-AB + C Circlotron Amplifier
Long-time readers know how far I have stretched the boundaries of Circlotron circuitry. They have seen class-G Circlotrons, hybrid Circlotrons, Solid-State Circlotrons, Circlotron & impedance-multiplier circuits, solid-state supercharged Circlotrons, and B+ supercharged Circlotrons. Well, now you can add class-AB + C Circlotrons to the list.

Back in 2009, blog number 177, I described my attempts at creating a constant-gm output stage as class-AC. (A better name would be "class AB + C.") But before diving into the new circuits, I want to mention how I was haunted by a previous class AB + C design effort, but I could not remember where I saw it. I had been corresponding with Bob Cordell about gm-doubling and I wanted find the earlier article. Google was of no help, so I went hunting through my vast collection of old magazines. I knew that I had seen the article while in high school, so at least I could narrow my search to the years between 1970 and 1974. It took me a while to find it, but there it was: the Edwin class A+B amplifier from the 3rd issue of 1972 Audio Amateur.

As I photo-scanned the page in the old magazine, I remembered my original discomfort with the illustration back in 1972. How so? I was convinced back then and I am today convinced that they got the schematic wrong, as the bottom diode is incorrectly placed. (I am sure that the Audio Amateur graphic artist had been given a sloppy schematic on a paper napkin and he (or she) couldn't see how the diodes must be connected.) The output stage drawing should have looked like this:

Now, the output stage makes sense—sort of; I believe the diodes were added only to stress the the point that the class-B transistors were not connected to the load at idle. In contrast, the two class-A transistors never cut off, while the two class-B only turn on as they are needed to fill in for the class-A pair. Actually, that put it the other way around, as Edwin's aim was to run two class-B output transistors (effectively class-C, wherein both devices are cut off at idle) and then use a pair of class-A transistors to fill in the void, wherein both of the class-B transistors didn't conduct. Therefore, his class-A transistors ran in very wimpy class-A mode, as were were only needed to fill in a small hole. As for the diodes, they are not needed and were not used in his actual amplifiers, example-1  and example-2. Here is another example from the Audio Amateur article.

Such an arrangement will still create discontinuities in the transfer curve; two of them, in fact, two where each of the class-B transistors kick in, as these transistors do not offer the same output impedance that the class-A transistors do through their emitter resistors.

On the other hand, if we run the first pair of transistors in class-AB, and set up the turn-on timing, the phasing of the class-C transistors perfectly, we can achieve a constant gm output stage. For example, with dead short to ground, this following output stage exhibits a constant gm (transconductance) of about 2A per input volt. With an 8-ohm load, the effective gm falls because of the 8-ohm load, which limits the current flow to about 1V/8.5-ohms. In other words, the higher the load impedance, the lower the effective gm of the output stage. With no load, changes in input voltage to not result in current-flow changes in the output stage. In terms of output impedance, the above output stage presents about 0.5 ohms of impedance. Of course, with a big dollop of negative feedback, the Zo will fall even lower. Still, 0.5 ohms isn't too bad, (a damping factor of 16) and is far lower than the typical tube amplifier's output impedance.

In the above circuit, the assumption is that the class-C transistors require a base-to-emitter voltage of 0.6V to turn on. Your mileage may vary, as they say. With MOSFETs, the turn-on voltage will be much higher, say between 1V to 5V; with vacuum tubes, the turn-on voltage will be negative, roughly equal to Vp/-mu.

One key point to my efforts at a constant-gm output stage, besides the benefits that accrue from a constant gm, is that it allows us to run much heavier idle current through the class-AB devices, be they tubes, MOSFETs, or transistors. In contrast, a typical solid-state power amplifier that uses bipolar transistors usually run 40mA to 50mA at idle. The big problem is maintaining that trivial idle-current flow, as the transistor's base-to-emitter voltage drops as the transistor heats up, which requires elaborate and finicky circuitry to counter the change in transistors voltage effectively.

In contrast, my 300mA of idle current (and the base voltages) is so fat that it can absorb a lot of slop and still work well, as the big 1-ohm emitter resistors are excellent current limiters that prevent the output stage from melting down, should the speaker cables short. But what of the high output impedance, resulting from such high-ohmage emitter resistors? If multiple pairing of class-AB+C transistors were used, as three pairs per channel, then 3-ohm emitter resistors could be used, which would definitely provide a wide safety margin.

The following images are from my blog number 177 post, which you should read.

Transistors Q1 & Q3 draw current at idle, but transistors Q2 & Q4 do not. The net result is a constant gm, which effectively equals twice that of a single device.

Once again, a better name is class-AB + C, as transistors Q1 & Q3 do cutoff, so they do not run in class-A.

Okay, John, this is all wonderful, but where are the dang tubes?

There coming. Imagine if we replace the class-AB transistors with triodes, so the triodes will run in class-AB, while the remaining transistors will run in class-C.

In other words, at idle and within the class-A envelope of current output, only the tubes will drive the speaker. But once we leave class-A, and one of the triodes ceases to conduct current, its counter-paired afterburner transistor fires up, drawing the make up current needed to yield a constant-gm output stage.  With such an arrangement, we could build either of the following OTL output stages.

The design on the left offers the advantage of using only NPN transistors, but the disadvantage of extremely high heat dissipation from the transistors when they are engaged. The design on the right offers much more sane dissipation requirements from the transistors, but uses PNP and NPN transistors. Both designs present the typical problems that all push-pull totem-pole topologies face of dissimilar drive signals and the need to use feedback in the creation of the drive signals to ensure the same output impedance from top and bottom devices.

In contrast, this arrangement within a Circlotron design, allows us to use only NPN types and to present equal-amplitude drive signals to both triodes.

Here we see the class AB+C Circlotron power amplifier's naked output stage. A total of eight triodes and eight NPN transistors are used. The triodes see a B+ voltage of 90V at idle; the transistors, 30V. But as the transistors are turned off at idle, they should run fairly cool. The reason the transistors do not get 4.7-ohm emitter resistors is to bring there transconductance up a bit after a small initial emitter-follower loss. The two crisscrossed -34V power-supply rails provide the needed negative bias voltage for the 6AS7 triodes. They are crisscrossed to better null DC offsets, as the crisscrossing produces a garter-belt biasing effect. If one side goes too positive, the other side sees the increase in bias voltage as prompt to increase it output voltage more positively, while the too negative side's negative drift forces the other side to move away from being too positive. Very simple DC feedback that works well. Here is how it would be actually implemented.

Note how -60V power-supply rails are used. Do not panic and falsely imagine that many power transformers are required. They are not. We can use just two 44Vac CT windings per channel with the following power supply circuit.

Sneaky, no? What is not shown is the heater power supply, which could consist of one massive 6.3Vac power transformer. Such a brute would have to deliver 2.5A of current per 6AS7 used. The sneakier approach would be to use DC to heat the 6AS7 heater elements. What? Yes, DC. If eight 6AS7 tubes (16 triodes) per channel, then we could string four heaters in series and use the 30Vdc power-supply rails to power the heater strings through a 2-ohm resistor. Thus, each channel would hold two heater strings, each holding four tube heaters.

By the way, we could go all-solid-state and use expensive and fancy lateral MOSFETs, such as the BUZ901.

With such a pairing, the power supply becomes simple indeed, as two 30V floating rails are needed per channel.  In the above example, the MOSFET idles at 300mA and the NPN transistor is cut off at idle. Once the MOSFET draws over 600mA, its partner transistor turns on, creating a constant-gm output. Why not use two MOSFETs? we certainly could.

Note the 40V power-supply rails, which would allow at least 60W of output into 8-ohm loads. Also note the relatively cheap IRF250 MOSFETs. The only problem with using MOSFETs throughout is getting the timing (the phasing of the turn on and turn off of the output devices) just right, as they do not have as fixed a turn-on voltage as NPN transistors do. Still, as the IRF250 is so cheap, we could buy a bunch and hand match them, which would make things a bit easier for us.

Note how the afterburner MOSFET overlap slightly with the primary class-AB MOSFETs. By the way, here is what just two IRF250 MOSFETs in the Circlotron look like.

A peak voltage output of 24Vpk equals 36W (rms) into 8-ohm loads, which implies 3Apk of current flow, which the graph shows not to be the case. Interesting. I must have run the SPICE again with the same drive-voltage settings as in the -2gm version. In other words, this version is getting a small break. Nonetheless, note the gm doubling, where the two MOSFETs overlap in conduction. (The idle current was a truly hot 0.5A, so class-A operation extended out to 1A and 4W output.) Let's look at the distortion harmonics now.

I know which I would prefer to listen to, as the lower 3rd harmonic on the anti-2gm Circlotron with four output devices is much lower. By the way, some argue that MOSFETs are exempt from gm-doubling distortion. Well, the above graph belies such a conclusion.

Okay, back to tubes. The two 30V power-supply rails limit the maximum output wattage into 8-ohm loads to about 40W. If more 6AS7 and MJE15032 pairs were used along with higher power-supply rail voltage, say 40Vdc to 55Vdc, then we could get much more power.

 

 

Class A + AB + C Circlotron Amplifier
Okay, fasten your mental seatbelts, as we are about to go on quite a ride. What if we run the triodes in strict class-A, so they never completely turn off? What if we run some transistors in class-AB, so that they are on at idle and turn off when the output swings big in the other direction? What if we add some extra transistors that run in class-C, which means that they are off at idle and only turn on when they are needed to create a constant-gm output? Well, we would get something like this:

Or, something like this, if we went all-solid-state.

The advantage that the hybrid version offers is that tubes look cool. The advantage the all-solid-state version offers is lower distortion. Even with with all the help provided by the added transistors, the 6AS7 is not all that linear to start with and the resulting distortion is still relatively high.

Here is the complete output stage.

Click on schematic to see enlargement

Only two 6AS7 are needed per channel (four triodes in other words). The following current plots from SPICE reveal that the 6AS7s never cutoff, even with 36W out into the 8-ohm load. (Green and red are the tube current plots.)

The above graph also shows how the class-AB and class-C transistors are timed correctly. The result is a constant-gm output stage.

Of course we could scale up, using more 6AS7s and transistors and higher rail voltages, which could deliver lower distortion and much higher output wattages. On the other hand, the following design is cheap, simple, and effective.

Click on schematic to see enlargement

The MOSFETs run in class-A and present a very high input impedance, which allows the simple DC servos to be used. The left DC servo sets the idle current, while the right DC servo sets the DC offset. The idle-current-setting DC servo only works because the MOSFET it controls runs in class-A. Had it run in class-AB, then the servo would over compensate during loud passages of music reproduction.

Next Time
Many new developments.

 

 

*
The exception is when such a two-gm device is used with a complementary driver stage that also uses a two-gm device, with both devices idling at the bend in their gm. In such a setup, the bends would cancel. Back in the the 1950s, the altogether amazing J. Ross MacDonald and Cathode Ray (writing in Wireless World) both pointed this possibility out.

 

    

User Guides for GlassWare Software
Just click on any of the above images to download a PDF of the user guides. By the way, all the links for the PCB user guides shown at the right now work.

Since I am still getting e-mail asking how to buy these GlassWare software programs: